Output phase modulation entrainment containment for digital filters

ABSTRACT

Method and apparatus for entrainment containment in digital filters using output phase modulation. Phase change is gradually introduced into the acoustic feedback canceller loop to avoid entrainment of the feedback canceller filter. Various embodiments employing different output phase modulation approaches are set forth and time and frequency domain examples are provided. Additional method and apparatus can be found in the specification and as provided by the attached claims and their equivalents.

RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.14/105,269, filed on 13 Dec. 2013, which application is a continuationof and claims the benefit of priority under 35 U.S.C. §120 to U.S.patent application Ser. No. 12/980,720, filed Dec. 29, 2010, whichapplication is a divisional of and claims the benefit of priority under35 U.S.C. §120 to U.S. patent application Ser. No. 11/276,763, filed onMar. 13, 2006, which applications are hereby incorporated by referenceherein in their entirety.

TECHNICAL FIELD

This disclosure relates to digital filters used for feedback and echocancellation, and in particular to method and apparatus for digitalfilters employing entrainment containment.

BACKGROUND

A feedback canceller or an echo canceller is a system that eliminates,as much as possible, the output of a system from re-entering its input.In the case of an audio system having a microphone, an audio processingunit and a speaker (or any other audio transducer), it is known that theoutput signal can leave the speaker and come back to the inputmicrophone by means of a physical acoustic path. This physical pathwhere the output sound waves can propagate back to the input of thesystem is usually referred as “acoustic feedback path”.

The reentrant signal can be perceived as an echo if the feedback pathdelay is long and the gain is low. This is usually the case oftelecommunication systems such as speakerphones. The system used toreduce such artifact is usually known as echo canceller.

If the feedback delay is shorter, but the loop gain is greater than one,it may cause a sustained oscillation. This is usually the case ofhearing aids and amplified mic/headset pairs. It is perceived as a loudwhistle, which forces the user to remove the apparatus from his/herears. It can also be perceived as a ringing artifact if the oscillationgets attenuated. The system used to reduce such artifacts is usuallyknown as feedback canceller.

Both echo canceller and feedback canceller are usually implemented as anadaptive system, whose goal is to match the system response of theacoustic feedback path. If the acoustic feedback path can be estimated,the feedback signal can also be estimated by supplying it with theoutput of the system.

When an acoustic feedback control system is stimulated with a sinusoidalsignal from the environment in an adaptive digital filter, the adaptivealgorithm will correlate the output of the filter with the feedbacksignal and with the stimulus signal itself. This will cause a degradedresponse to the feedback signal. This phenomenon is called “entrainment”as the feedback canceller gets entrained by the stimulus signal. Ithappens with signals that have high autocorrelation between samples,such as sinusoidal signals and other periodic signals.

The entrainment causes several effects upon the performance of thefeedback canceller:

-   -   a. Entrainment degrades the estimate of the feedback signal,        because its response gets distracted to the auto-correlated        signal input. Therefore, the system has decreased feedback        cancellation.    -   b. Entrainment causes attenuation of the input stimulus signal.    -   c. Entrainment increases the instability of the system. Once the        periodic input stimulus is removed, the entrained system might        immediately act as a feedback generator itself, which can cause        sustained oscillation. This condition can get worse the longer        the periodic input signal is allowed to stimulate the system, as        the coefficients of some filter designs can grow indefinitely.    -   d. Entrainment degrades the response of a longer digital filter.        Under entrainment, the longer the digital filter, the worse its        response because the smaller coefficients at the tail of the        filter are more sensitive to get mistuned by the entraining        input signal.

What is needed in the art is an improved system for avoiding orcontaining entrainment of digital filter designs. The system should bestraightforward to implement in a variety of applications.

SUMMARY

The above-mentioned problems and others not expressly discussed hereinare addressed by the present subject matter and will be understood byreading and studying this specification.

The present subject matter provides method and apparatus for entrainmentcontainment of digital filter systems. The present subject matterrelates to time domain and frequency domain embodiments for entrainmentcontainment of digital filter systems. Several embodiments are providedwhich relate to digital filters for acoustic feedback reduction. Someapplications include hearing assistance devices, such as hearing aids.

For example, one such apparatus includes a sensor to receive sound andconvert it to an electrical signal; an analog-to-digital converter toconvert the electrical signal into a digital signal; a summing nodereceiving the digital signal and an acoustic feedback compensationsignal Y(z) adapted to correct for acoustic feedback received by thesensor, the summing node providing an error signal E(z) by subtractionof the acoustic feedback compensation signal Y(z) from the digitalsignal; a signal processing module and a phase adjustment moduleprocessing the error signal E(z) in series to produce an output signalX(z); an adaptive filter including an adaptive algorithm receiving theerror signal E(z), the adaptive filter producing the feedbackcompensation signal Y(z); a digital-to-analog converter providing ananalog version of the output signal X(z); and a receiver to outputprocessed sound from the analog version, wherein the phase adjustmentmodule gradually changes phase applied to the output signal X(z).Various embodiments including phase adjustment modules which graduallychanges phase between zero and 180 degrees are provided. One such moduleincludes an all-pass filter. Various phase shift increments areperformed from about 0.25 degrees to 25 degrees. Embodiments of 0.25 and4, and 25 degree increments are some examples. Some examples includeprogrammable phase changes.

Another example provided has a phase adjustment module which graduallychanges phase between zero and 360 degrees. One such example is a pairof all-pass filters. Various phase shift increments are performed fromabout 0.25 degrees to 25 degrees. Embodiments of 0.25 and 4, and 25degree increments are some examples. Some examples include programmablephase changes.

Frequency domain embodiments are also provided. Some applicationsinclude hearing assistance devices, such as hearing aids. One suchapplication includes a sensor to receive sound and convert it to anelectrical signal; an analog-to-digital converter to convert theelectrical signal into a digital signal; a frequency analysis module toproduce frequency domain subband signals from the digital signal; asumming node receiving the frequency domain subband signals and acousticfeedback compensation subband signals AFC(k) adapted to correct foracoustic feedback received by the sensor, the summing node providingerror subband signals E(k) by subtraction of the received signals; acomplex multiplier producing a gradually phase shifted version of theerror subband signals E′(k); a signal processing module to process thephase shifted error subband signals E′(k); a time synthesis module toproduce time domain, processed, digital signals; a digital-to-analogconverter providing an analog version of the processed, digital signals;and a receiver to produce processed sound from the analog version;wherein the processed, digital signals are passed through a bulk delayand converted back into frequency domain signals to be used for acousticfeedback cancellation by an adaptive filter which produces the acousticfeedback compensation subband signals AFC(k).

Various phase adjustments may be made, for example, in some embodimentsthe multiplier receives a gradually shifted phase signal from a phaseshifter, and the phase shifter module resets when reaching an aggregatephase of 360 degrees. Various phase increments include 4 degrees, 0.25degrees, 25 degrees, or any phase between about 0.25 degrees to about 25degrees. Various time-domain to frequency domain transformations can beused, including FFT, and its inverse, the IFFT, can be used to get backinto the time domain. In some examples the time and/or frequencyanalysis modules include a weighted overlap-add structure. Differentadaptive filters designs may be used, such as an LMS adaptive filterdesign.

Methods for entrainment containment are also provided. One method forentrainment containment, includes converting an analog sound signal intoa digital signal; processing the digital signal using an acousticfeedback reduction loop; gradually changing phase of a forward feed ofthe feedback reduction loop; converting the processed digital signalsinto analog signals; and generating sound from the analog time domainsignals.

Different phase changes are possible. In one example, phase is changedat about 4 degree increments per sample. In one example, phase ischanged at about 0.25 to about 25 degree increments per sample. Invarious applications, larger phase shift increments are used to achievemore aggressive entrainment containment. In various applications,smaller phase shift increments are used to reduce artifacts.

Some frequency domain methods include converting analog sound signalsinto a plurality of digital, frequency domain subband signals;processing a gradually phase shifted version of the digital, frequencydomain subband signals to create processed digital, frequency domainsignals to reduce entrainment; converting the processed digital,frequency domain signals into analog time domain signals; and generatingsound from the analog time domain signals. Some applications includegradually incrementing phase of the digital, frequency domain subbandsignals at 4 degree increments per sample. Some applications includegradually incrementing phase of the digital, frequency domain subbandsignals at about 0.25 to about 25 degree increments per sample. In someapplications, larger phase shift increments are used to achieve moreaggressive entrainment containment. In some applications, smaller phaseshift increments are used to reduce artifacts.

This Summary is an overview of some of the teachings of the presentapplication and not intended to be an exclusive or exhaustive treatmentof the present subject matter. Further details about the present subjectmatter are found in the detailed description and appended claims. Otheraspects will be apparent to persons skilled in the art upon reading andunderstanding the following detailed description and viewing thedrawings that form a part thereof, each of which are not to be taken ina limiting sense. The scope of the present invention is defined by theappended claims and their legal equivalents.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows one example of signal processing in a time-domain outputphase modulator approach, according to one embodiment of the presentsubject matter.

FIG. 2 shows one example of an all-pass filter for use in a time-domainoutput phase modulator approach, according to one embodiment of thepresent subject matter.

FIG. 3 shows one example of an all-pass filter for use in a time-domainoutput phase modulator approach, according to one embodiment of thepresent subject matter.

FIG. 4 shows a mapping of a variable q to variable a, for use to controlan all-pass filter for use in a time-domain output phase modulatorapproach, according to one embodiment of the present subject matter.

FIG. 5 shows one example of a series of all-pass filters for use in atime-domain output phase modulator approach, according to one embodimentof the present subject matter.

FIGS. 6, 8, and 10 demonstrate entrainment of a hearing assistancedevice upon reception of room noise and a 500 Hz tone and with theoutput phase modulator deactivated.

FIGS. 7, 9, and 11 show a reduction in entrainment of an output phasemodulator operating on the same systems as FIGS. 6, 8 and 10,respectively.

FIG. 12 is a frequency domain approach to containing entrainment usingoutput phase modulation, according to one embodiment of the presentsubject matter.

DETAILED DESCRIPTION

The following detailed description of the present subject matter refersto subject matter in the accompanying drawings which show, by way ofillustration, specific aspects and embodiments in which the presentsubject matter may be practiced. These embodiments are described insufficient detail to enable those skilled in the art to practice thepresent subject matter. References to “an”, “one”, or “various”embodiments in this disclosure are not necessarily to the sameembodiment, and such references contemplate more than one embodiment.The following detailed description is demonstrative and not to be takenin a limiting sense. The scope of the present subject matter is definedby the appended claims, along with the full scope of legal equivalentsto which such claims are entitled.

The present subject matter relates to methods and apparatus forentrainment containment in digital filter designs. The principlesprovided by this disclosure may be applied in the time domain or in thefrequency domain. They may be applied in a variety of apparatus,including, but not limited to hearing assistance devices. Although theexamples set forth herein relate to hearing assistance devices, those ofskill in the art will understand other applications and variationsfalling within the scope of the present subject matter upon reading andunderstanding this specification. The method and apparatus set forthherein are demonstrative of the principles of the invention, and it isunderstood that other method and apparatus are possible using theprinciples described herein.

Time Domain Examples

FIG. 1 demonstrates one example of an acoustic feedback cancellationsystem using a novel output phase modulation according to one embodimentof the present subject matter. This system 100 can be adapted for use inaudio applications, including but not limited to, hearing assistancedevices. One application involving hearing assistance devices is the useof hearing aids. In such an application, the sound sensor 101 is amicrophone or other acoustic sensor for receiving sound and convertingit into electrical signals. Sensor 101 receives sound via the acousticfeedback path 130 and other sounds from outside the system. Acousticfeedback path 130 represents sound from the audio source 112 thatreaches sensor 101, thus making a closed loop feedback of acousticsound. The audio source 112 is a receiver (also known as a speaker) forcreating sound based on electrical signals presented to it. Applicationsbesides hearing assistance devices are possible which employ theprinciples set forth herein.

The AD block 102 of system 100 converts an analog input signal into adigital output signal. In various embodiments, the AD block 102 includesan analog-to-digital converter and may include various amplifiers orbuffers to interface with sensor 101. Digital signals representing thesuperposition of acoustic feedback and other sounds are processed by theclosed loop system of block 120.

The DA block 111 converts the incoming digital signal into an analogoutput signal. In various embodiments, the DA block 111 includes andigital-to-analog converter and may include various amplifiers or signalconditioners for conditioning the analog signal for the audio source112. In hearing assistance devices, such as hearing aids, the audiosource is called a “receiver.” In other applications the audio sourcemay be a speaker or other sound transducer.

Block 120 represents a simplified flow of the digital signal processingof input signals from AD block 102. In one embodiment, block 120 isimplemented using a digital signal processor (DSP) for echocancellation/feedback cancellation in the digital domain. The filter 108is used to emulate the acoustic feedback path 130 in the digital domain.In various embodiments, an adaptive algorithm, such as an LMS algorithm(least mean squares algorithm) 106 is used to tune the filter 108response such that it matches the acoustic feedback path 130 response.The estimated feedback signal, Y(z), 109 can then be generated byapplying the output of the system, X(z), 107 to the filter 108. Theestimated feedback signal Y(z) 109 is subtracted from the input signalin the digital domain using summer 105, therefore reducing the effectsof the acoustic feedback path 130.

Module 104 includes various different types of signal processing thatthe system may employ. For example, in cases where the signal processingis for a hearing assistance device, module 104 is adjusted for the besthearing of the wearer of the device. In cases where the hearingassistance device is a hearing aid, module 104 provides hearing aidsignal processing. Such processing is known to include adjustments ofgain and phase for the benefit of the hearing aid user.

In one embodiment, filter 108 is a FIR filter (finite impulse responsefilter). Thus, an adaptive algorithm, such as the LMS algorithm is usedto tune the FIR coefficients based on the correlation of the input errorsignal E(z) 113 and the output of the system X(z) 107. In variousembodiments, a bulk delay line is inserted between the output of thesystem and the FIR input, if the FIR is not long enough to accommodatethe feedback path length, therefore being truncated. In one embodiment,filter 108 is an IIR filter (infinite impulse response filter). Otherfilters may be used without departing from the scope of the presentsubject matter.

It is understood that the LMS algorithm is not the only adaptivealgorithm that can be used. Other such algorithms include, but are notlimited to N-LMS and filtered X-LMS algorithms. The N-LMS algorithm is avariation of the LMS algorithm which also uses the power of both E(z)and X(z) signals to adjust the tuning step of the FIR filter, which isbased on the correlation of the same two signals. The filtered-X LMSalgorithm is a variation of the LMS algorithm which uses filtered X(z)and E(z) samples to generate the correlation to tune the filter 108.

Output phase modulator 110 is used to adjust the phase of the output ofmodule 104 in the feedback loop to contain entrainment of the filter108. It was discovered that by controlled adjustment of phase of theoutput signal X(z), entrainment of the filter 108 can be reduced oravoided. The process can avoid coefficient drift caused by entrainmentand can correct coefficients that have drifted due to an onset ofentrainment. Various embodiments will be provided herein to show howphase may be adjusted to avoid entrainment.

Output Phase Modulator Using 180 Degree Switching

One embodiment of the output phase modulator switches phase 180 degreesevery half period of a periodic cycle time, T. One way to do this isusing the following approach:

 sample_count = 0  For every new output sample out(n)  if (sample_count< T/2)    out(n) = out(n) // Don't do anything  else    out(n) = −out(n)// Reverse the output, 180 degrees out of phase    if (sample_count=T)    sample_count=0     endif  endif   sample_count = sample_count + 1

By this approach, the same reversed output signal is applied to thefilter 108 as well as the acoustic feedback path 130. This causes thefilter 108 coefficients to keep the same correlation to the acousticfeedback path response, but it reverses the correlation of the samecoefficients to the periodic input stimulus. The reversed correlation tothe periodic input stimulus will cause the coefficients to move to theopposite direction, therefore canceling the previous entrainment drift.In this way, the entrainment effect is contained by making thecoefficients to move back and forth around the correct values, insteadof allowing them to drift to improper (entrained) values.

The output phase reversal in its basic version as described aboveillustrates its working principles, but, because of the abrupt change itcauses in the phase of the signal, it generates audible artifacts.

Another implementation of the output phase modulator 110 looks foropportunities to change the phase, where it will not cause artifacts,thereby being unperceivable to the user. These windows of opportunityhappen when there is “notch” in the signal power envelope. A notchdetector constantly monitors the power envelope of the signal, and whenit detects a notch, it flags the system an opportunity to reverse thephase.

One implementation of the notch detector is to run two signal envelopedetectors, a slow one and a fast one. The magnitude of the slow envelopedetector is compared to the magnitude of the fast envelope detector bymeans of division (or division approximation). In a variation of thisembodiment, subtraction of the outputs of the two signal envelopedetectors may be used. The difference is a metric of the relationshipbetween the slow one and the fast one. Other comparisons may beperformed without departing from the scope of this subject matter.

If the slow envelope magnitude is bigger than the power of the fastenveloped magnitude by a certain threshold value, a notch is detected.

The system described above works well for inputs having notches, such asfor speech signals and music signals. But not as effectively on steady,constant amplitude sinusoidal inputs as they lack power envelopenotches. The next embodiment provides an approach to accommodatedifferent input signals.

Single Stage Output Phase Modulator

FIG. 2 shows a block diagram of a single stage approach to output phasemodulation employing an all-pass filter. In one embodiment, the singleall-pass filter is capable of different amounts of phase shift, and upto 180 degrees of phase shift. One such all-pass filter is shown in FIG.3. This embodiment uses a single coefficient, a, as a control of theoverall phase adjustment. As a is adjusted from +1 to −1, phase isshifted from 0 degrees to 180 degrees. The system is stable anddistortion free as long as a is varied in small increments. The transferfunction for the embodiment of FIG. 3 is:

H(z)=(1+α*z)/(z+α).

The relationship between α and phase shift caused by alpha is non-linearand described by the equation:

Phase shift=−2A TAN [((1−α)/(1+α))*TAN(M*θ/2)], where θ is frequency inradians.

Thus, a first order all-pass filter can be used to smoothly change thephase shift from 0 to 180 degrees by varying α from +1 to −1 and thenback from −1 to +1. The α increment (or step size) produces a nonlinearphase shift, and so the empirical equation for α is:

α=2.0156*(2̂q/2 ̂7)−1.0156,

where q varies from 7 to 0 in small steps.

FIG. 4 is a chart showing a nonlinear mapping of variable q to α. Oneexample of q varying in small steps is q varying by 0.001. Some examplesof q and α in the extreme are:

when q=7, α=1; and

when q=0, α=−1.

In various embodiments, the system smoothly changes the phase of theoutput: from 0 degrees to 180 degrees, and then from 180 degrees back to0 degrees. In some embodiments, this system smoothly sweeps the phaseshift from 0 to 180, going through several intermediate values (1, 2, 3. . . , 179, 180).

In various embodiments, different incremental changes in phase areemployed. In one embodiment, phase change increments of 0.25 degrees to25 degrees are programmable. In various embodiments, a fixed phasechange increment is employed. In one embodiment, a phase change of 4degree increments is used. In one embodiment, a phase change of 0.25degree increments is used. In one embodiment, a phase change of 25degree increments is used. The greater the phase change increment, thefaster entrainment is compensated for and the larger the audibleartifacts. Thus, smaller phase changes result in lower artifacts, but inslower compensation for entrainment. If transitions are made slowly, thetransitions cause no perceivable artifacts. Other embodiments arepossible without departing from the scope of the present subject matter.

Implementations using all-pass filters change the phase of the inputsignal without changing its magnitude. The amount of phase shift can becontrolled by slowly changing the coefficient(s) of the filter.

FIGS. 6 and 7 demonstrate efficacy of the single stage embodiment. Thesefigures show an output of a digital signal processing system of ahearing aid sampling at 16 KHz (simulation) receiving a 500 Hz tone todemonstrate the effects of the algorithm on entrainment. FIG. 6 showsthe hearing aid where the filter 108 is entrained by the 500 Hz tone.FIG. 7 shows what happens when the present algorithm is enabled, therebyeliminating entrainment.

Two-Stage Output Phase Modulator

FIG. 5 shows a two-stage series of all-pass filters to achieve a 360degree phase shift. The input is A(z) and the ultimate output is B(z).In varying embodiments, the first and second filters operate as follows:Both filters start off at zero degrees, then the first all-pass filterbegins sweeping from zero to 180 degrees. Once the first filter sweepsto 180 degrees, it stays there and the second filter begins sweepingfrom zero to 180 degrees. The aggregate phase shift of the seriesfilters is 360 degrees. Upon reaching 360 degrees the second all-passfilter begins sweeping back to zero degrees. Upon reaching zero degrees,the second all-pass filter stops sweeping and the first all-pass filterstarts sweeping from 180 degrees towards zero degrees. When both filtersare at zero degrees the process starts over again.

It is understood that any number of combinations of filter adjustmentsdesigned to provide unity gain and sweeps from zero to 360 degrees totalare provided by the disclosed structure. It is understood that the exactorder of sweeping can vary without departing from the scope of theprinciples set forth herein.

In various embodiments, different incremental changes in phase areemployed. In one embodiment, phase change increments of 0.25 degrees to25 degrees are programmable. In various embodiments, a fixed phasechange increment is employed. In one embodiment, a phase change of 4degree increments is used. In one embodiment, a phase change of 0.25degree increments is used. In one embodiment, a phase change of 25degree increments is used. The greater the phase change increment, thefaster entrainment is compensated for and the larger the audibleartifacts. Thus, smaller phase changes result in lower artifacts, but inslower compensation for entrainment. If transitions are made slowly, thetransitions cause no perceivable artifacts. Other embodiments arepossible without departing from the scope of the present subject matter.

The equations in the previous section for coefficient α are incorporatedhere. Now that two filters are used, the equations have an a1 and an a2(one for each stage). The following algorithm identifies the input ofthe first filter as x1 and the input of the second filter as y1 (theoutput of the first filter is y1). The output of the second filter isy2. The coefficients for phase change of the first filter are a1 and q1and the coefficients for the second filter are a2 and q2.

An algorithm as follows may be employed:

 if AlgoON==1   x1=out;  % Filter 1   y1=(x1−yold1)*a1 + xold1;  xold1=x1;   yold1=y1;  % Filter 2   y2=(y1−yold2)*a2 + xold2;  xold2=y1;   yold2=y2;   final_out=y2;  % Update alpha  % Find next q  if f1==1    q1=q1+inc1;    if (q1>=7)     q1−7;     inc1=−inc1;    f1=0;     f2=1;   elseif (q1<=0)     q1=0;     inc1=−inc1;     f1=0;    f2=1;    end   end   if f2==1    q2=q2+inc2;    if (q2>=7)     q2=7;    inc2=−inc2;     f2=0;     f1=1;    elseif (q2<=0)     q2=0;    inc2=−inc2;     f2=0;     f1=1;    end   end  % Find next alpha  a1=2.0156*(2{circumflex over ( )}q1/2{circumflex over ( )}7)−1.0156;  a2=2.0156*(2{circumflex over ( )}q2/2{circumflex over ( )}7)−1.0156;End

Thus, in the first part of a cycle, the first filter gradually changesthe phase of the signal from 0 to 180 degrees and the second filterremains static (phase shift at 0 degrees). In the second part of a cyclethe first filter now remains static at 180 degrees phase shift, and nowthe second filter sweeps from 0 to 180 degrees. In the third part of thecycle, the first filter sweeps from 180 degrees back to 0 degrees whilethe second filter remains static at 180 phase shift. In the last andfourth part of the cycle, the first filter remains static at 0 degrees,and the second filter sweeps from 180 degrees back to 0 degrees. Thewhole pattern repeats again for every T samples.

FIGS. 8 to 11 demonstrate efficacy of the two-stage embodiment. Thesefigures show an output of a digital signal processing system of ahearing aid sampling at 16 KHz (simulation) receiving a 500 Hz tone todemonstrate the effects of the algorithm on entrainment. FIG. 8 shows anenvelope of the output of the hearing aid without the present algorithm.In FIG. 8, it can be seen that extra frequency is introduced duringentrainment. FIG. 9 shows the effect of activating the algorithm, which,besides envelope modulation, presents only the sinusoid with no extrafrequency introduced. FIGS. 10 and 11 are before and after spectralgraphs of the output of the hearing aid without and with, respectively,the algorithm on. It is straightforward to see entrainment and extraharmonics output by the hearing aid without the present algorithmrunning on the hearing aid (FIG. 10). The entrainment is gone after thealgorithm is turned on (FIG. 11). Thus, efficacy of the approach isestablished.

Frequency Domain Examples

FIG. 12 shows one embodiment of an output phase modulation approach inthe frequency domain. This system 1200 can be adapted for use in audioapplications, including but not limited to, hearing assistance devices.One application involving hearing assistance devices is the use ofhearing aids. In such an application, the sound sensor 1201 is amicrophone or other acoustic sensor for receiving sound and convertingit into electrical signals. Sensor 1201 receives sound via the acousticfeedback path 1230 and other sounds from outside the system. Acousticfeedback path 1230 represents sound from the audio source 1212 thatreaches sensor 1201, thus making a closed loop feedback of acousticsound. The audio source 1212 is a receiver (also known as a speaker) forcreating sound based on electrical signals presented to it. Applicationsbesides hearing assistance devices are possible which employ theprinciples set forth herein.

The AD block 1202 of system 1200 converts an analog input signal into adigital output signal. In various embodiments, the AD block 1202includes an analog-to-digital converter and may include variousamplifiers or buffers to interface with sensor 1201. Digital signalsrepresenting the superposition of acoustic feedback and other sounds areprocessed by the closed loop system 1200.

The DA block 1211 converts the incoming digital signal into an analogoutput signal. In various embodiments, the DA block 1211 includes adigital-to-analog converter and may include various amplifiers or signalconditioners for conditioning the analog signal for the audio source1212. In hearing assistance devices, such as hearing aids, the audiosource is called a “receiver.” In other applications the audio sourcemay be a speaker or other sound transducer.

FIG. 12 represents a simplified flow of the digital signal processing ofsignals from sensor 1201 to audio source 1212. The “T” inputs to variousmodules indicate that such operations are synchronous in one embodiment.In one embodiment, the processing is implemented using a digital signalprocessor (DSP) for echo cancellation/feedback cancellation in thedigital domain. In the present frequency based approach, the frequencyanalysis modules 1222 and 1218 convert digital, time domain signals intofrequency subband signals (subband signals denoted with a “(k)” toindicate that the signal is subdivided into frequency bands forprocessing). Time synthesis module 1216 converts the subband frequencydomain signals into time domain signals. One such approach forconversion includes, but is not limited to, the use of weighted overlapstructures for discrete Fourier transforms (DFTs), such as thosediscussed in Multirate Digital Signal Processing, by Ronald E. Crochiereand Lawrence R. Rabiner, Prentice-Hall, 1983, especially at Section7.2.5, starting on p. 313, the entire book hereby incorporated byreference. One such approach is a fast Fourier transform (FFT) forconversion to the frequency domain and an inverse FFT or IFFT forconversion to the time domain. Other conversion method and apparatus maybe employed without departing from the scope of the present subjectmatter.

The filter 1208 is used to emulate the acoustic feedback path 1230 inthe frequency subband digital domain. In various embodiments, anadaptive algorithm, such as a sub-band LMS algorithm (least mean squaresalgorithm) 1206 is used to tune the filter 1208 response such that itmatches the acoustic feedback path 1230 response. The estimated feedbacksignal, AFC(k) 1209, can then be generated by applying a delayed versionof the output of the system, X(z) 1207, to the filter 1208. The bulkdelay 1220 provides a time domain delay to X(z) 1207, before convertingthe signals back into the subband frequency domain using frequencyanalysis module 1218. The estimated feedback signal, AFC(k) 1209, issubtracted from the input signal m(k) in the subband frequency domainusing summer 1205, therefore reducing the effects of the acousticfeedback path 1230.

Module 1204 includes various different types of subband frequency domainsignal processing that the system may employ. For example, in caseswhere the signal processing is for a hearing assistance device, module1204 is adjusted for the best hearing of the wearer of the device. Incases where the hearing assistance device is a hearing aid, module 1204provides hearing aid signal processing. Such processing is known toinclude adjustments of gain and phase for the benefit of the hearing aiduser.

In one embodiment, filter 1208 is a FIR filter (finite impulse responsefilter). Thus, an adaptive algorithm, such as the LMS algorithm is usedto tune the FIR coefficients based on the correlation of the input errorsignal E(k) 1213 and the delayed output of the system X(z) 1207 which isconverted to the frequency domain, S_(d)(k). The bulk delay 1220 isbetween the output of the system and the FIR input, so that the FIR islong enough to accommodate the feedback path length, without beingtruncated.

It is understood that the LMS algorithm is not the only adaptivealgorithm that can be used. Other such algorithms include, but are notlimited to N-LMS and filtered X-LMS algorithms. The N-LMS algorithm is avariation of the LMS algorithm which also uses the power of both E(k)and S_(d)(k) signals to adjust the tuning step of the FIR filter, whichis based on the correlation of the same two signals. The filtered-X LMSalgorithm is a variation of the LMS algorithm which uses filteredS_(d)(k) and E(k) samples to generate the correlation to tune the filter1208.

Output phase modulator is comprised of phase shifter 1214 and a complexmultiplier 1210. The combination is used to adjust the phase of theinput of module 1204 in the feedback loop to contain entrainment of thefilter 1208. It was discovered that by controlled adjustment of phase ofthe output signal E′(k) 1217, entrainment of the filter 1208 can bereduced or avoided. The process can avoid coefficient drift caused byentrainment and can correct coefficients that have drifted due to anonset of entrainment. Various embodiments will be provided herein toshow how phase may be adjusted to avoid entrainment. In one embodiment,phase shifter 1214 increments phase by a predetermined amount and cyclesfrom 0 to 360 degrees, then starts over again at 0 degrees andincrements to 360 degrees.

In various embodiments, different incremental changes in phase areemployed. In one embodiment, phase change increments of 0.25 degrees to25 degrees are programmable. In various embodiments, a fixed phasechange increment is employed. In one embodiment, a phase change of 4degree increments is used. In one embodiment, a phase change of 0.25degree increments is used. In one embodiment, a phase change of 25degree increments is used. The greater the phase change increment, thefaster entrainment is compensated for and the larger the audibleartifacts. Thus, smaller phase changes result in lower artifacts, but inslower compensation for entrainment. If transitions are made slowly (forexample, around 1 Hz or less), the transitions cause no perceivableartifacts. Other embodiments are possible without departing from thescope of the present subject matter.

Frequency Domain Enhancements

Various enhancements can be made to the frequency domain embodiments setforth herein. For example, to reduce artifacts, the output phasemodulation can be disabled for low frequencies. For example, theprocessing of certain subbands below a predetermined frequency thresholdcan disable the output phase modulator, since entrainment is generallynot an issue at lower frequencies. In one embodiment, output phasemodulation is disabled for frequencies below 1250 Hz. Various frequencythresholds may be used without departing from the scope of the presentsubject matter. This approach also avoids phase change artifacts whichare more noticeable at low frequencies.

Another enhancement to the previous threshold frequency approach is tohave a transition band where output phase modulation is optionaldepending on the energy detected about the frequency of the threshold.This reduces or eliminates audible artifacts arising from phasediscontinuities due to switching the device output phase modulation onand off at the threshold frequency. One approach to performing thetransition is to window the energy just below the threshold frequency(i.e., window the energy from about 750 Hz to 1250 Hz). As energyincreases in the window, program the system to disable output phasemodulation at all frequencies and freeze the adaptive filter. As energydecreases, enable the output phase modulation and start adapting thefilter again. In one embodiment, a window energy for frequencies between750 Hz and 1250 Hz is detected and compared with energy found in theother bands. If the energy in the window is greater than the energy inthe other bands by 12 dB, then output phase modulation is disabled andadaptation is stopped. If the energy in all bands is less than athreshold energy, then output phase modulation is also disabled andadaptation is stopped.

Other enhancements are possible without departing from the scope of thepresent subject matter.

The output phase modulation system (OPM), for containing the entrainmenteffect was presented above. Various embodiments are provided whichreduce the phase reversal artifact to unperceivable levels. The proposedembodiments reduce the effect of entrainment upon the filter 108coefficients and corrects coefficient entrainment drift. The presentsubject matter improves feedback signal estimation, therefore improvingthe feedback cancellation. It avoids the attenuation of the inputsignal. It causes increased system stability by removal of the periodicinput signal, the feedback canceller won't act as a feedback generatoras before. It avoids the indefinite growth of the coefficients, anothercause of system instability when the system is being constantlystimulated by a periodic signal. It also allows longer filter (filter108 for the frequency domain approach, and/or filter 1208 for thefrequency domain approach) to be used; the effect of entrainment uponthe smaller tail coefficient is reduced. Other benefits are also enjoyedwhich are not enumerated expressly herein.

These principles apply not only to the feedback canceller, but also tothe echo canceller described above. Other applications may benefit usingthe present principles set forth herein.

It is understood that various hardware, firmware, and softwarerealizations are possible without departing from the scope of thepresent subject matter. Variations are also possible which do not departfrom the present teachings. For instance, if a signal processor includedanalog to digital conversion electronics, it is understood that FIG. 1,blocks 120 and 102 could be realized by one signal processor. If asignal processor included a digital to analog conversion, then blocks120 and 111 may be realized by a single processor. Likewise, in FIG. 12,it is possible that any combination of blocks could be realized by asingle processor. For example, if a signal processor included analog todigital conversion, then it could include A/D converter 1202 with theremaining portion of the system. If a signal processor included digitalto analog conversion, then the system could be realized in a singleprocessor which would perform the system functions and that of D/Aconverter 1211. Thus, the examples set forth here are intended todemonstrate the principles of the present subject matter, but are notintended to be exclusive or exhaustive of the many variations andrealizations possible.

It is further understood that the principles set forth herein can beapplied to a variety of hearing assistance devices, including, but notlimited to occluding and non-occluding applications. Some types ofhearing assistance devices which may benefit from the principles setforth herein include, but are not limited to, behind-the-ear devices,over-the-ear devices, on-the-ear devices, and in-the ear devices, suchas in-the-canal and/or completely-in-the-canal hearing assistancedevices. Other applications beyond those listed herein are contemplatedas well.

CONCLUSION

This application is intended to cover adaptations or variations of thepresent subject matter. It is to be understood that the abovedescription is intended to be illustrative, and not restrictive. Thus,the scope of the present subject matter is determined by the appendedclaims and their legal equivalents.

What is claimed is:
 1. An apparatus, comprising: a sensor to receivesound and convert it to an electrical signal; an analog-to-digitalconverter to convert the electrical signal into a digital signal; afrequency analysis module to produce frequency domain subband signalsfrom the digital signal; a summing node receiving the frequency domainsubband signals and acoustic feedback compensation subband signalsAFC(k) adapted to correct for acoustic feedback received by the sensor,the summing node providing error subband signals E(k) by subtraction ofthe received signals; a complex multiplier producing a gradually phaseshifted version of the error subband signals E′(k); a hearing aid signalprocessing module to process the phase shifted error subband signalsE′(k); a time synthesis module to produce time domain, processed,digital signals; a digital-to-analog converter providing an analogversion of the processed, digital signals; and a receiver to produceprocessed sound from the analog version; wherein the processed, digitalsignals are passed through a bulk delay and converted back intofrequency domain signals to be used for acoustic feedback cancellationby an adaptive filter which produces the acoustic feedback compensationsubband signals AFC(k).
 2. The apparatus of claim 1, wherein themultiplier receives a gradually shifted phase signal from a phaseshifter, and the phase shifter module resets when reaching an aggregatephase of 360 degrees.
 3. The apparatus of claim 2, wherein the phaseshifter shifts phase in 4 degree increments.
 4. The apparatus of claim2, wherein the phase shifter changes phase in 0.25 degree phaseincrements.
 5. The apparatus of claim 2, wherein the phase shifterchanges phase in 25 degree phase increments.
 6. The apparatus of claim2, wherein the phase shifter is programmable to change phase inincrements varying from between about 0.25 degrees to about 25 degrees.7. The apparatus of claim 1, wherein the frequency analysis moduleincludes an FFT.
 8. The apparatus of claim 1, wherein the time synthesismodule includes an IFFT.
 9. The apparatus of claim 1, wherein thefrequency analysis module includes a weighted overlap-add structure. 10.The apparatus of claim 1, wherein the time synthesis module includes aweighted overlap-add structure.
 11. The apparatus of claim 1, whereinthe adaptive filter includes a subband LMS adaptive filter design.
 12. Amethod for entrainment containment, comprising: converting analog soundsignals into a plurality of digital, frequency domain subband signals;processing a gradually phase shifted version of the digital, frequencydomain subband signals to create processed digital, frequency domainsignals to reduce entrainment; converting the processed digital,frequency domain signals into analog time domain signals; and generatingsound from the analog time domain signals.
 13. The method of claim 12,further comprising gradually incrementing phase of the digital,frequency domain subband signals at 4 degree increments per sample. 14.The method of claim 12, further comprising gradually incrementing phaseof the digital, frequency domain subband signals at about 0.25 to about25 degree increments per sample.
 15. The method of claim 12, whereinlarger phase shift increments are used to achieve more aggressiveentrainment containment.
 16. The method of claim 12, wherein smallerphase shift increments are used to reduce artifacts.